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LT3587 High Voltage Monolithic Inverter and Dual Boost FEATURES
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DESCRIPTION
The LT(R)3587 provides a one chip solution for applications requiring two positive and one negative high voltage supplies. The LT3587 input voltage range of 2.5V to 6V makes it ideal for various battery-powered systems. A single resistor programs each of the three output voltage levels and the output current of Boost3. The intelligent softstart allows for sequential soft-start of the Boost1 output followed by the negative output with a single capacitor. Internal sequencing circuitry also disables the inverter until the Boost1 output has reached 87% of its final value. The LT3587 integrates all the power switches, soft-start, and output-disconnect circuits into a small 3mm x 3mm QFN package. This high level of integration combined with small external components makes the LT3587 ideal for space constrained applications.
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.
Ideal for CCD, LCD, LED Backlight and OLED Applications Easy Generation of 15V (50mA), -8V (100mA) and 20V (20mA) from a Li-Ion Cell VVIN Range: 2.5V to 6V Wide Output Ranges: Up to 32V for the Boosts and Up to -32V for the Inverter Output Disconnect for the Boost Channels Boost3 Allows Voltage Programming and/or Current Programming for a `One Wire Current Source' Overload Fault Protection with Fault I/O Pin Indicator Combined Soft-Start and Enable Pins Small 20-Pin 3mm x 3mm QFN Package
APPLICATIONS
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Digital Still and Video Cameras Scanner and Display Systems PDA, Cellular Phones and Handheld Computers LED Backlight and OLED Display Drivers CCD Imager Bias General High Voltage Supply Bias
TYPICAL APPLICATION
Li-Ion Powered Supply for CCD Imager and Six White Backlight LEDs
VVIN 2.5V TO 6V 1F 2.2F SW3 CAP3 IFB3 8.06k GND LT3587 EN/SS1 EN/SS3 FLT SW2 2.2F VVIN 2.5V TO 6V 15H 15H 22F CCD NEGATIVE -8V, 100mA
3587 TA01a
10H
15H 90 VIN SW1 CAP1 VOUT1 FB1 10F CCD POSITIVE 15V, 50mA EFFICIENCY (%) 1M 2.7pF LED DRIVER 20mA, UP TO 6 LEDS 1M 6.8pF
Efficiency Curve
CDD+ = 50mA - 85 CDD = 100mA LED = 20mA 80 75 70 65 60 55 50 0 0.5 1 NORMALIZED CURRENT ALL CHANNELS ENABLED CDD- LED POWER DISSIPATION ALL CHANNELS ENABLED CDD+ 800 700 POWER DISSIPATION (mW) 600 500 400 300 200 100 0 1.5
3587 TA01b
VOUT3 VFB3 FB2
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LT3587 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW EN/SS3 EN/SS1 15 FB1 14 VOUT1 21 13 CAP1 12 GND 11 SW1 6 FB2 7 GND 8 GND 9 10 SW2 NC VFB3 IFB3 VIN
VIN ..............................................................................6V Soft-Start Input Pins EN/SS1, EN/SS3 .....................................................6V Feedback Pins FB1, FB2, IFB3, VFB3 ................................. -0.2V to 6V High Voltage Switch Pins SW1, SW2, SW3...................................................40V High Voltage Output Pins CAP1, CAP3, VOUT1, VOUT3 ...................................32V Bidirectional I/O Pin FLT ..........................................................................6V FLT Current ........................................................10mA Operating Junction Temperature Range .. -40C to 125C Storage Temperature Range................... -65C to 125C
20 19 18 17 16 VOUT3 1 CAP3 2 SW3 3 GND 4 FLT 5
UD PACKAGE 20-LEAD (3mm 3mm) PLASTIC QFN JA = 68C/W, JC = 4.2C/W EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LT3587EUD#PBF TAPE AND REEL LT3587EUD#TRPBF PART MARKING LDNC PACKAGE DESCRIPTION 20-Lead (3mm x 3mm) Plastic QFN TEMPERATURE RANGE -40C to 125C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VVIN = 3.6V, VEN/SS1 = VEN/SS3 = VVIN unless otherwise noted (Note 2, 3).
PARAMETER Operating Input Voltage Range Quiescent Current VEN/SS1 = 0V, VEN/SS3 = VVIN OR VEN/SS1 = VVIN, VEN/SS3 = 0V OR VEN/SS1 = VEN/SS3 = VVIN Not Switching VEN/SS1 = VEN/SS3 = 0V, In Shutdown Switching Frequency Maximum Duty Cycle Minimum On Time Power Fault Delay from Any Output to FLT FLT Input Threshold Low FLT Leakage Current FLT Voltage Output Low VFLT = 5V IFLT = 1mA
l l l l l
ELECTRICAL CHARACTERISTICS
CONDITIONS
MIN 2.5
TYP 2.4
MAX 6 4
UNITS V mA
5.5 0.8 87 1 93 50 16 0.4 1.0
9 1.2 70 1.6 1 0.4
A MHz % ns ms V A V
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LT3587 ELECTRICAL CHARACTERISTICS
PARAMETER Boost1 CAP1 Bias Current FB1 Reference Voltage VOUT1 Output Voltage SW1 Current Limit SW1 VCESAT SW1 Leakage Current EN/SS1 for Full Inductor Current EN/SS1 Shutdown Voltage Threshold EN/SS1 Pin Bias Current VOUT1 Current Limit CAP1 to VOUT1 On-Resistance (RDISC1) VOUT1 Disconnect Leakage Inverter FB2 Reference Voltage Output Voltage SW2 Current Limit SW2 VCESAT SW2 Leakage Current FB1 Threshold to Start Negative Channel Boost3 CAP3 Bias Current Boost3 Programmed Current VFB3 Reference Voltage VOUT3 Output Voltage SW3 Current Limit SW3 VCESAT SW3 Leakage Current EN/SS3 for Full Inductor Current EN/SS3 Shutdown Voltage Threshold EN/SS3 Pin Bias Current VOUT3 Current Limit CAP3 to VOUT3 On Resistance (RDISC3) VOUT3 Disconnect Leakage CAP3 Pin Overvoltage Clamp Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. VEN/SS3 = 0V VCAP3 = 15V, VIFB3 = 0V VCAP3 = 15V, IVOUT3 = 20mA VVIN = VCAP3 = 6V, VVOUT3 = 0V 27 ISW3 = 200mA VSW3 = 15V VVFB3 = VIFB3 = 0.6V
l l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VVIN = 3.6V, VEN/SS1 = VEN/SS3 = VVIN unless otherwise noted (Note 2, 3).
CONDITIONS VCAP1 = 15V, VOUT1 = Open
l
MIN
TYP 60
MAX 150 1.25 15.75
UNITS A V V mA mV
1.19 14.25 800
1.22 15 990 200 0.1
RFB1 = 1M ISW1 = 400mA VSW1 = 15V VFB1 = 1.1V, VFB2 = 0.1V
l
5 2.5
A V V A mA A mV V mA mV
l
0.2 -0.5 100 -1 155 5 0.1 -10 -7.5 900 5 -8 1090 250 0.1 87 70 5 90 150 22 0.83 16 8 1 20 -8.5 -1.5
VEN/SS1 = 0V VCAP1 = 15V VCAP1 = 15V, IVOUT1= 50mA VVIN = VCAP1 = 6V, VVOUT1 = 0V
RFB2 = 1M ISW2 = 600mA VSW2 = 15V Percent of Final Regulation Value VCAP3 = 15V, VOUT3 = Open RIFB3 = 8.06k RVFB3 = 1M, IVOUT3 = 20mA RVFB3 = 1M, IVOUT3 = 20mA
l
A % A mA V V mA mV
l l l
18 0.77 14 400
20 0.8 15 480 250 0.1
5 2
A V V A mA A V
0.2 -0.5 70 -1 110 10 0.1 29 15 1 31 -1.5
Note 2: All currents into pins are positive; all voltages are referenced to GND unless otherwise noted. Note 3: The LT3587 is guaranteed to meet specified performance from 0C to 125C. Specifications over the -40C to 125C operating range are assured by design, characterization and correlation with statistical process controls.
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LT3587 TYPICAL PERFORMANCE CHARACTERISTICS Specifications are at TA = 25C unless otherwise noted.
Shutdown Quiescent Current vs Input Supply Voltage
10 VVIN = 3.6V EN/SS1 = EN/SS3 = 0V QUIESCENT CURRENT (A) 8 90C 6 25C 125C QUIESCENT CURRENT (mA) 3.0 3.5 VVIN = EN/SS1 = EN/SS3 = 3.6V FB1 = VFB3 = 1.5V FB2 = IFB3 = 0V 125C 90C UVLO VOLTAGE (V) 2.15
Quiescent Current When On But Not Switching vs Input Supply Voltage
2.20
VIN UVLO Voltage vs Temperature
2.5
25C
2.10
4
-40C
2.0
-40C
2.05
2 2.5
3
3.5 4 5 4.5 INPUT VOLTAGE (V)
5.5
6
1.5 2.5
3
3.5 4 5 4.5 INPUT VOLTAGE (V)
5.5
6
2.00 -50
-25
0 25 75 50 TEMPERATURE (C)
100
125
3587 G01
3587 G02
3587 G03
FB1, VFB3 and IFB3 Regulation Voltage vs Temperature
1.240 FB2 REGULATION VOLTAGE (mV) VVIN = 3.6V RFB1 = RVFB3 = 1M RIFB3 = 8.06k 1.230 IVOUT1 = 50mA IVOUT3 = 20mA 0.825 VIFB3, VVFB3 REGULATION VOLTAGE (V) 15
FB2 Regulation Voltage vs Temperature
VVIN = 3.6V RFB2 = 1M IVNEG = 100mA
REGULATION VOLTAGE (V)
10
VFB3
0.813
5
1.220 FB1 1.210
VIFB3
0.800
0
0.788
-5
1.200 -50
-25
0 25 75 50 TEMPERATURE (C)
100
0.775 125
-10 -50
-25
0 25 75 50 TEMPERATURE (C)
100
125
3587 G04
3587 G05
FB1, VFB3, IFB3 Bias Current in Regulation vs Temperature
14.50 14.25 BIAS CURRENT (A) 14.00 13.75 13.50 IIFB3 IVFB3 106 104 IFB3 BIAS CURRENT (A) 102 100 98 96 94 125 FB2 BIAS CURRENT (A) -7.75
FB2 Bias Current in Regulation vs Temperature
VVIN = 3.6V RFB2 =1M IVNEG = 100mA
-7.85
IFB1
-7.95
-8.05
VVIN = 3.6V RFB1 = RVFB3 =1M 13.25 RIFB3 =8.06k IVOUT1 = 50mA IVOUT3 = 20mA 13.00 -50 -25 50 0 25 75 TEMPERATURE (C)
-8.15
100
-8.25 -50
-25
0 25 75 50 TEMPERATURE (C)
100
125
3587 G06
3587 G07
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LT3587 TYPICAL PERFORMANCE CHARACTERISTICS
Switching Frequency vs Temperature
1.20 0.5 VVIN = 3.6V 0.4 1.10 FREQUENCY (MHz) VCESAT (V) 0.3 SW1 CURRENT LIMIT (A) SW3 SW2
Specifications are at TA = 25C unless otherwise noted. Switches Current Limit vs Temperature
1.2 1.0 0.8 0.6 0.4 0.2 0 -50 SW3 SW1 SW2 VVIN = 3.6V DUTY CYCLE = 60%
Switches VCESAT vs Current
VVIN = 3.6V
1.00
0.2
0.90
0.1
0.80 -50
0 -25 0 25 75 50 TEMPERATURE (C) 100 125
0
0.2
0.6 0.4 0.8 SW CURRENT (A)
1
1.2
3587 G09
-25
0 25 75 50 TEMPERATURE (C)
100
125
3587 G08
3587 G10
Switches Current Limit vs Duty Cycle
2.5 VVIN = 3.6V 2.0 CURRENT LIMIT (A) CURRENT LIMIT (A) SW2 1.5 SW1 1.2 1.6
SW1 and SW2 Current Limit vs EN/SS1 Voltage
0.5 VVIN = 3.6V DUTY CYCLE = 60% SW2 0.4 CURRENT LIMIT (A)
SW3 Current Limit vs EN/SS3 Voltage
VVIN = 3.6V SW3
SW1 0.8
0.3
1.0 SW3
0.2
0.4
0.5
0.1
0
0
10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%)
3587 G11
0 0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 EN/SS1 VOLTAGE (V)
3587 G12
0 0.5 0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 EN/SS3 VOLTAGE (V)
3587 G13
Output Disconnects On Resistance vs Temperature (RDISC1, RDISC3)
VVIN = 3.6V IVOUT1 = 50mA 12 IVOUT3 = 20mA ON RESISTANCE () 10 8 6 4 2 -50 RDISC1 RDISC3 OUTPUT DISCONNECTS CURRENT LIMIT (mA) 14 200.0
Output Disconnects Current Limit vs Temperature
VVIN = 3.6V VVOUT1 = VVOUT3 = 15V PULL-UP CURRENT (A) 1.50
EN/SS1, EN/SS3 Pull-Up Current In Shutdown vs Temperature
VVIN = 3.6V EN/SS1 = EN/SS3 = 0V
167.5 IVOUT1 135.0
1.25
1.00
102.5 IVOUT3 70 -50
0.75
-25
50 0 25 75 TEMPERATURE (C)
100
125
-25
50 0 25 75 TEMPERATURE (C)
100
125
0.50 -50
-25
50 0 25 75 TEMPERATURE (C)
100
125
3587 G14
3587 G15
3587 G16
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LT3587 TYPICAL PERFORMANCE CHARACTERISTICS Specifications are at TA = 25C unless otherwise noted.
EN/SS1, EN/SS3 Shutdown Threshold vs Temperature
0.45 SHUTDOWN THRESHOLD VOLTAGE (V) VVIN = 3.6V CAP3 OV CLAMP VOLTAGE (V) 31
CAP3 Overvoltage Clamp vs Temperature
VVIN = 3.6V
0.40
30
0.35
29
0.30
0.25
28
0.20 -50
-25
0 25 75 50 TEMPERATURE (C)
100
125
27 -50
-25
50 0 25 75 TEMPERATURE (C)
100
125
3587 G17
3587 G18
PIN FUNCTIONS
VOUT3 (Pin 1): Boost3 Output Pin. This pin is the drain of an output disconnect PMOS transistor. CAP3 (Pin 2): Boost3 Output Capacitor Pin. This pin is the source of an output PMOS disconnect. Connect a capacitor from this pin to ground. SW3 (Pin 3): Boost3 Switch Pin. Connect an inductor from this pin to VIN. Minimize trace area at this pin to minimize EMI. GND (Pin 4, 7, 8, 12): Ground Pins. FLT (Pin 5): Fault Pin. This pin is a bidirectional opendrain pull-down pin. This pin pulls low when any of the enabled outputs fall out of regulation for more than 16ms. Each output is ignored during start-up until its respective enable/soft-start pin allows for full inductor current. This pin can also be externally forced low to disable all the supply outputs. Once this pin goes low (either due to an out of regulation condition or externally forced low), the pin latches low until the inputs to EN/SS1 and EN/SS3 are set low or the input supply pin is recycled. Pull up this pin to VIN with a 200k resistor when not used. FB2 (Pin 6): Inverter Output Voltage Feedback Pin. Connect a resistor RFB2 from this pin to the Inverter Output (VNEG) such that: RFB2 = |VNEG|/8A Note that FB2 pin voltage is about 0V when in regulation. There is an internal 153k resistor from the FB2 pin to the internal reference. SW2 (Pin 9): Inverter Switch Pin. Connect an inductor between this pin and VIN, as well as the flying capacitor from this pin to the anode of the lnverter ground return diode. Minimize trace area at this pin to minimize EMI. NC (Pin 10): No Connect Pin. Leave open or connect to ground. SW1 (Pin 11): Boost1 Switch Pin. Connect an inductor from this pin to VIN. Minimize trace area at this pin to minimize EMI. CAP1 (Pin 13): Boost1 Output Capacitor Pin. This pin is the source of an output PMOS disconnect. Connect a capacitor from this pin to ground.
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LT3587 PIN FUNCTIONS
VOUT1 (Pin 14): Boost1 Output Pin. This pin is the drain of an output disconnect PMOS transistor. FB1 (Pin 15): Boost1 Output Voltage Feedback Pin. Connect a resistor RFB1 from this pin to VOUT1 (or CAP1) such that: RFB1 = ((VVOUT1/1.22V) - 1) * 88.5k There is an internal 88.5k resistor from the FB1 pin to ground. EN/SS1 (Pin 16): Boost1/Inverter Shutdown and Soft-Start Pin. Boost1 and Inverter are enabled when the voltage on this pin is greater than 2.5V. They are disabled when the voltage is below 0.2V. An internal 1A current source in conjunction with an external capacitor can be used to ramp this pin and provide soft-start. VIN (Pin 17): Input Supply Pin. Must be locally bypassed with an X5R or X7R type ceramic capacitor. EN/SS3 (Pin 18): Boost3 Shutdown and Soft-Start Pin. Boost3 is enabled when the voltage on this pin is greater than 2V. It is disabled when the voltage is below 0.2V. An internal 1A current source in conjunction with an external capacitor can be used to ramp this pin and provide soft-start. IFB3 (Pin 19): Boost3 Output Current Programming Pin. Connect a resistor RIFB3 from this pin to ground such that: RIFB3 = 200 * (0.8V/IVOUT3) If Boost3 output is configured as a voltage regulator, RIFB3 can be optionally used to limit the maximum output current to ILIMIT: RIFB3 = 200 * (0.8V/ILIMIT) Note: Tie IFB3 to GND when no current limit is desired. VFB3 (Pin 20): Boost3 Output Voltage Feedback Pin. Connect a resistor RVFB3 from this pin to VOUT3 (or CAP3) such that: RVFB3 = ((VVOUT3/0.8V) - 1) * 56.3k There is an internal 56.3k resistor from the VFB3 pin to ground. In the current regulator configuration, RVFB3 can be optionally used to limit the maximum output voltage to VCLAMP, such that: RVFB3 = ((VCLAMP/0.8V) - 1) * 56.3k Note: When no voltage clamp is desired in the current regulator configuration, tie VFB3 to GND. Exposed Pad (Pin 21): Ground Pin. Connect to PCB ground plane. Ground plane connection through multiple vias under the package is recommended for optimum electrical and thermal performance.
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LT3587 BLOCK DIAGRAM
VIN EN/SS3 C5 200mV - FLT VOUT3 RVFB3 VFB3 VMAX 56.3k RAMP GENERATOR SHDN3 A8 1A L4 SW3 SOFTSTART A5 VC3 VIN
VIN FLT EN/SS3 C6 PTAT BIAS EN/SS1 EN/SS3 OSCILLATOR VC1 VC2 VC3 FILTER AND 16ms DELAY R S Q L1 SW1 DS1 CAP1 X1 A3 R S Q Q1 M1 VOUT1 DISCONNECT CONTROL C1 FLT VIN RIFB3 100k
BANDGAP AND LDO VOUT1 RFB1 FB1
88.5k FLT
200mV
- +
A7
VREF
1.22V
EN/SS1 C3
1A VIN SOFTSTART
153k FB2 L2
VNEG GND
8
+
RFB2
-
+ -
A1
+
VREF 0.8V
OVERVOLTAGE PROTECTION
DS3 CAP3 C4 VOUT3
-
A6 R
X3 S Q Q3 M2 M3
+
DISCONNECT CONTROL
IFB3
VIN
EN/SS1 VC1
- +
-
+
RAMP GENERATOR SEQUENCING A2 VC2
SHDN1
VIN
SW2
-
A4 R
X2 S Q Q2 C2
+
RAMP GENERATOR
DS2
L3 VNEG C7
3587 F01
Figure 1. Block Diagram
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LT3587 OPERATION
All three channels of the LT3587 use a constant frequency, current mode control scheme to provide voltage and/or current regulation at the output. Operation can be best understood by referring to the Block Diagram in Figure 1. If EN/SS1 is pulled higher than 200mV, the bandgap reference, the start-up bias and the oscillator are turned on. At the start of each oscillator cycle, the SR latch X1 is set, which turns on the power switch Q1. A voltage proportional to the switch current is added to a stabilizing ramp and the resulting sum is fed into the positive terminal of the PWM comparator A3. When this voltage exceeds the level at the negative input of A3, the SR latch X1 is reset, turning off the power switch Q1. The level at the negative input of A3 is set by the error amplifier A1, which is simply an amplified version of the difference between the reference voltage of 1.22V and the feedback voltage. In this manner, the error amplifier sets the correct peak switch current level to keep the output voltage in regulation. If the error amplifier output increases, more current is delivered to the output; if it decreases, less current is delivered. The second channel is an inverting converter. This channel is also enabled through the EN/SS1 pin. The basic operation of this second channel is the same as the positive channel. The SR latch X2 is also set at the start of each oscillator cycle. The power switch Q2 is turned on at the same time as Q1. Q2 turns off based on its own feedback loop, which consists of error amplifier A2 and PWM comparator A4. The reference voltage of this negative channel is ground. Voltage clamps (not shown) on the output of the error amplifiers A1 and A2 enforce current limit on Q1 and Q2 respectively. Similar to the first channel, the third channel is also a positive boost regulator. If EN/SS3 is pulled higher than 200mV, the bandgap reference, the start-up bias and the oscillators are also turned on. The SR latch X3 is set at the start of each oscillator cycle which turns on the power switch Q3. Q3 turns off based on its own feedback loop, which consists of error amplifier A5 and PWM comparator A6. The level at the negative input of A6 is set by the error amplifier A5, and is an amplified version of the difference between the reference voltage of 0.8V and the maximum of the two feedback voltages at VFB3 and IFB3. A separate comparator (not shown) sets the maximum current limit on Q3. The IFB3 pin is pulled up internally with a current that is (1/200) times the load current out of the VOUT3 pin. Therefore, an external resistor connected from this pin to ground generates a feedback voltage proportional to the VOUT3 output load current at the IFB3 pin. When the voltage at VFB3 is higher than the voltage at IFB3, the third channel regulates to the feedback voltage at VFB3, which in normal application is a divided down voltage from VOUT3. In this state, the third channel behaves as a boost voltage regulator. On the other hand if the voltage at IFB3 is higher, the third channel regulates to the feedback voltage at IFB3, which therefore regulates the VOUT3 output load current to a particular value. In this state, the third channel behaves as a boost current regulator. PMOS M1 is used as an output disconnect pass transistor for the first channel. M1 disconnects the load (VOUT1) from the input as long as the voltage between CAP1 and VIN is less than 2.5V (typical) and the voltage between CAP1 and VOUT1 is less than 10V (typ). Similarly, PMOS M3 is used as an output disconnect pass transistor for the third channel. M3 disconnects the load (VOUT3) from the input when the third channel is in shutdown (EN/SS3 voltage is lower than 200mV) and the voltage between CAP3 and VOUT3 is less than 10V (typical).
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LT3587 APPLICATIONS INFORMATION
Inductor Selection A 15H inductor and a 10H inductor are recommended for the LT3587 Boost1 channel and Boost3 channel respectively. The inverting channel can use 15H or 22H inductors. Although small size is the major concern for most applications, for high efficiency the inductors should have low core losses at 1MHz and low DCR (copper wire resistance). The inductor DCR should be on the order of half of the switch on resistance for its channel: 0.5 for Boost1, 0.4 for the inverter and 1 for Boost3. For robust applications, the inductors should have current ratings corresponding to their respective peak current during regulation. Furthermore, with no soft-start, the inductor should also be able to withstand temporary high start-up currents of 1A, 1.1A and 480mA for the Boost1, inverter and Boost3 channels respectively (typ, refer to the Typical Performance Characteristics curves). Capacitor Selection The small size of ceramic capacitors makes them suitable for LT3587 applications. X5R and X7R types of ceramic capacitors are recommended because they retain their capacitance over wider voltage and temperature ranges than other types such as Y5V or Z5U. A 1F input capacitor is sufficient for most LT3587 applications. The output capacitors required for stability depend on the application. For most applications, the output capacitor values required are: 10F for the Boost1 channel, 22F for the inverter channel and 2.2F for the Boost3 channel. The inverter requires a 2.2F flying capacitor. Note that this flying capacitor needs a voltage rating of at least VIN + |VNEG|. Inrush Current When a supply voltage is abruptly applied to the VIN pin, the voltage difference between the VIN pin and the CAP pins generates inrush current. For the case of the Boost1 channel, the inrush current flows from the input through the inductor L1 and the Schottky DS1 to charge the Boost1 output capacitor C1. Similarly for the Boost3 channel, the inrush current flows from the input through the inductor L4 and the Schottky DS3 to charge the output capacitor C4. For the inverting channel, the inrush current flows from the input through inductor L2, charging the flying capacitor C2 and returning through the Schottky diode DS2. The selection of inductor and capacitor values should ensure that the peak inrush current is below the rated momentary maximum current of the Schottky diodes. The peak inrush current can be estimated as follows: (VVIN - 0.6) * e L C 4L R2C -1
-1 -1 tan ()
IP =
=
where L is the inductance, C is the capacitance and R is the total series resistance in the inrush current path, which includes the resistance of the inductor and the Schottky diode. Note that in this equation, we model the Schottky as having a fixed 0.6V drop. Table 1 gives inrush peak currents for some component selections. Note that inrush current is not a concern if the input voltage rises slowly.
Table 1. Inrush Peak Current
VVIN (V) 5 5 5 3.6 3.6 3.6 R () 0.68 0.68 0.68 0.745 0.745 0.745 L (H) 15 22 10 15 22 10 C (F) 10 2.2 2.2 10 2.2 2.2 IP (A) 2.48 1.19 1.64 1.64 0.80 1.10
Schottky Diode Selection For any of the external diode (DS1, DS2 and DS3) selections, besides having sufficiently high reverse breakdown voltage to withstand the output voltage, both forward voltage drop and diode capacitance need to be considered. Schottky diodes rated for higher current usually have lower forward voltage drops and larger capacitance. Although lower forward voltage drop is good for efficiency, a large
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LT3587 APPLICATIONS INFORMATION
capacitance will slow down the switching waveform, which can cause significant switching losses at 1MHz switching frequency. Some recommended Schottky diodes are listed in Table 2.
Table 2. Recommended Schottky Diodes
DIODE FORWARD FORWARD CAPACICURRENT VOLTAGE TANCE (mA) DROP (V) (pF at 10V) MANUFACTURER 1000 500 500 500 500 520 0.35 0.49 0.43 0.48 0.48 0.53 30 25 8 39 20 17 ROHM www.rohm.com NXP/Phillips www.nxp.com Vishay www.vishay.com Diodes Inc. www.diodes.com Zetex www.zetex.com
The same constraints as the other Schottky diodes apply for selecting D3. Therefore, the same recommended Schottky diodes in Table 2 can be used for D3. Boost3 Overcurrent and Overvoltage Protection As briefly discussed in the Operation section, the regulation loop of Boost3 uses the maximum of the two voltages at VFB3 and IFB3 as feedback information to set the peak current of its power switch Q3. This allows for the Boost3 loop to be configured as either a boost voltage regulator or a boost current regulator (Figure 3). Furthermore, this architecture also allows for a programmable current limit on voltage regulation or voltage limit on current regulation.
VVIN VVIN
PART NUMBER RSX051VA-30 PMEG401OCEJ PMEG2005EB IR05H40CSPTR B0540WS ZLLS400
VIN
SW3 CAP3
VIN
SW3
Smaller Footprint Inverter Topology In certain applications with higher tolerance of current ripple at the output of the inverter, the inductor L3 can be replaced with a Schottky diode. Since the Schottky diode footprint is usually smaller than the inductor footprint, this alternate topology is recommended if a smaller overall solution is a must. Note that this topology is only viable if the absolute value of the inverter output is greater than VIN. This Schottky diode is configured with the anode connected to the output of the inverter and the cathode to the output end of the flying capacitor C2 as shown in Figure 2.
RFB1 1M FB2
3587 F02
LT3587 BOOST3 VOUT3 VOLTAGE RVFB3 REGULATOR VFB3 VOLTAGE IFB3 EN/SS3 REGULATION FEEDBACK RIFB3 RESISTOR
CAP3 LT3587 BOOST3 VOUT3 CURRENT RVFB3 REGULATOR VFB3 OPTIONAL IFB3 EN/SS3 PROGRAMMABLE VOLTAGE LIMIT RIFB3 RESISTOR CURRENT REGULATION FEEDBACK RESISTOR
3587 F03
OPTIONALPROGRAMMABLE CURRENT LIMIT RESISTOR
Figure 3. Boost3 Configured as a Voltage Regulator and as a Current Regulator
When configured as a boost voltage regulator, a feedback resistor from the output pin VOUT3 to the VFB3 pin sets the voltage level at VOUT3 at a fixed level. In this case, the IFB3 pin can either be grounded if no current limiting is desired or connected to ground with a resistor such that: ILIMIT = 200 * (0.8V/RIFB3) where ILIMIT is the desired output current limit value. Recall that the pull-up current on the IFB3 pin is controlled to be typically 1/200 of the output load current at the VOUT3 pin. In this case, when the load current is less than ILIMIT, the Boost3 loop regulates the voltage at the VFB3 pin to 0.8V. When there is an increase in load current beyond ILIMIT, the voltage at VFB3 starts to drop and the voltage at IFB3 rises above 0.8V. The Boost3 loop then regulates the voltage at the IFB3 pin to 0.8V, limiting the output
LT3587 SW2 L2 15H C2 2.2F
D3
VVIN 2.5V TO 4.5V
INVERTER OUTPUT -8V, 100mA
DS2
C7 22F
Figure 2. Inverter Configured with a Schottky Diode in Place of the Output Inductor
3587fc
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LT3587 APPLICATIONS INFORMATION
current at VOUT3 to ILIMIT. Figure 4 compares the transient responses with and without current limit when a current overload occurs.
VVOUT3 5V/DIV IVOUT3 13mA/DIV 15V
lower than 29V is obtained by connecting a resistor from the VOUT3 pin to the VFB3 pin such that: RFB3 = ((VCLAMP/0.8V) - 1) * 56.3k where VCLAMP is the desired output voltage clamp level. In this case, when the voltage level is less than VCLAMP, the Boost3 loop regulates the voltage at the IFB3 pin to 0.8V. When the output load fails open-circuit or is disconnected, the voltage at IFB3 drops to reflect the lower output current and the voltage at VFB3 starts to rise. When the voltage at VOUT3 rises to VCLAMP, the Boost3 loop then regulates the voltage at the VFB3 pin to 0.8V, limiting the voltage level at VOUT3 to VCLAMP. Figure 5 contrasts the transient responses with and without programmed VCLAMP when the output load is disconnected.
20mA LOAD STEP
IL4 200mA/DIV
3587 F04a 200s/DIV VVIN = 3.6V WITHOUT CURRENT LIMIT: IFB3 CONNECTED TO GND VOUT3 STAYS AT 15V, OUTPUT CURRENT INCREASES FROM 20mA TO 40mA
VVOUT3 5V/DIV IVOUT3 13mA/DIV
15V
VVOUT3 10V/DIV
20V OUTPUT LOAD DISCONNECTED
20mA LOAD STEP IL4 200mA/DIV
IL4 200mA/DIV
VVIN = 3.6V WITH 20mA CURRENT LIMIT: RIFB3 = 8.06k OUTPUT CURRENT STAYS AT 20mA, VOUT3 DROPS FROM 15V TO 7.5V
200s/DIV
3587 F04b
3587 F05a 200s/DIV VVIN = 3.6V WITHOUT PROGRAMMED OUTPUT VOLTAGE CLAMP: VFB3 CONNECTED TO GND
Figure 4. Boost3 Waveform in an Output Current Overload Event with and Without Output Current Limit
VVOUT3 10V/DIV
20V OUTPUT LOAD DISCONNECTED
The LT3587 CAP3 pin has an internal overvoltage protection. When the voltage at the CAP3 pin is driven above 29V (typ), the Boost3 loop is disabled and the SW3 pin stops switching. When configured as a boost current regulator, a feedback resistor from the IFB3 pin to ground sets the output current at VOUT3 at a fixed level. In this case, if the VFB3 pin is grounded then the overvoltage protection defaults to the open-circuit clamp voltage level of 29V. A voltage clamp
IL4 200mA/DIV
200s/DIV VVIN = 3.6V WITH PROGRAMMED OUTPUT VOLTAGE CLAMP AT 24V
3587 F05b
Figure 5. Boost3 Output Open-Circuit Waveform with and Without Programmed Output Voltage Clamp
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LT3587 APPLICATIONS INFORMATION
Setting The Output Voltages and The Boost3 Output Current The LT3587 has a trimmed internal feedback resistor. A 1M feedback resistor from each output pin to its corresponding feedback pin sets the outputs to 15V for Boost1, -8V for the inverter and 15V for Boost3. Note that only one resistor is needed to set the output voltage for each channel. Set the output voltages according to the following formulas: RFB1 = ((VVOUT1/1.22V) - 1) * 88.5k RFB2 = |VNEG|/8A RVFB3 = ((VVOUT3 /0.8V) - 1) * 56.3k As described in previous sections, Boost3 can be configured as a boost current regulator. When configured as such, set the output current according to the following formula: RIFB3 = 200 * (0.8V/IVOUT3) In order to maintain accuracy, use high precision resistors when setting any of the channels output voltage and/or the Boost3 output current (1% is recommended). Soft-Start The LT3587 has two soft-start control pins: EN/SS1 and EN/SS3. The EN/SS1 pin controls the soft-start for both the Boost1 and the inverter, while the EN/SS3 pin controls the soft-start for the Boost3. Each of these soft-start pins is pulled up internally with a 1A current source. Connecting a capacitor from the EN/SS1 pin to ground programs a soft-start ramp for the Boost1 and the inverter channels. Use an open-drain transistor to pull this pin low to shut down both the Boost1 and the inverter. Turning off this transistor allows the 1A pull-up current to charge the soft-start capacitor. When the voltage at the EN/SS1 pins goes above 200mV, the regulation loops for Boost1 and the inverter are enabled. The VC1 node voltage follows the EN/SS1 voltage as it continues to ramp up to ensure slow start-up on the Boost1 channel. The VC2 node follows the ramp voltage minus 0.7V. This ensures that the inverter starts up after the Boost1, but still has a slow ramping output to avoid large start-up currents. The Boost1 and the inverter regulation loops are free running with full inductor current when the voltage at the EN/SS1 pin is above 2.5V. Start Sequencing The LT3587 also has internal sequencing circuitry that inhibits the inverter channel from operating until the feedback voltage of the Boost1 voltage (at the FB1 pin) reaches about 1.1V (87% of the final voltage). This ensures that the Boost1 output voltage is near regulation before any negative voltage is generated at the inverter output. Figure 6 contrasts the start-up sequencing without any soft-start capacitor, and with a 10nF soft-start capacitor. Connecting a capacitor from the EN/SS3 pin to ground sets up a soft-start ramp for the Boost3 channel. As the 1A current charges up the capacitor, the Boost3 regulation loop is enabled when the EN/SS3 pin voltage goes above 200mV. The VC3 node voltage follows the EN/SS3 voltage as it ramps up ensuring slow start-up on the Boost3 channel. When the voltage at the EN/SS3 pin is above 2V, the Boost3 regulation loop is free running with full inductor current.
VEN/SS1 2V/DIV IVIN 500mA/DIV VVOUT1 10V/DIV VNEG 10V/DIV
0V
0mA
0V 0V
400s/DIV
3587 F06a
VEN/SS1 2V/DIV IVIN 500mA/DIV VVOUT1 10V/DIV VNEG 10V/DIV 0mA
0V 0V
4ms/DIV
3587 F06b
Figure 6. VEN/SS1, VOUT1, VNEG, IVIN with No Soft-Start Capacitor, and with a 10nF Soft-Start Capacitor
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LT3587 APPLICATIONS INFORMATION
Output Disconnect Both the Boost1 and the Boost3 channels have an output disconnect between their respective CAP pin and VOUT pin. This disconnect feature prevents a DC path from VIN to VOUT . For Boost1, this output disconnect feature is implemented using a PMOS (M1) as shown in the Block Diagram in Figure 1. When turned on, M1 is driven hard in the linear region to reduce power dissipation when delivering current between the CAP1 pin and the VOUT1 pin. M1 stays on as long as the voltage difference between CAP1 and VIN is greater than 2.5V. This allows for the positive bias to stay high for as long as possible as the negative bias discharges during turn off. The disconnect transistor M1 is current limited to provide a maximum output current of 155mA (typ). However, there is also a protection circuit for M1 that limits the voltage drop across CAP1 and VOUT1 to about 10V. When the voltage at CAP1 is greater than 10V, in an overload or a short-circuit event, M1 current is limited to 155mA until the voltage across CAP1 to VOUT1 grows to about 10V. Then M1 is turned on hard without any current limit to allow for the voltage on CAP1 to discharge as fast as possible. When the voltage across CAP1 and VOUT1 reduces to less than 10V, the output current is then again limited to 155mA. Figure 7 shows the output voltage and current during an overload event with VCAP1 initially at 15V. The output disconnect feature on Boost3 is implemented similarly using M3. However, in this case M3 is only turned off when the EN/SS3 pin voltage is less than 200mV and the Boost3 regulation loop is disabled. The disconnect transistor M3 is also current limited, providing a maximum output current at VOUT3 of 110mA (typ). M3 also has a similar protection circuit as M1 that limits the voltage drop across CAP3 and VOUT3 to about 10V. Figure 8 shows the output voltage and current during an overload event with VCAP3 initially at 24V.
IVOUT3 500mA/DIV 0mA IL4 500mA/DIV VCAP3 10V/DIV VVOUT3 10V/DIV 40s/DIV
3587 F08a
24V
IVOUT3 500mA/DIV 0mA IL4 500mA/DIV VCAP3 10V/DIV VVOUT3 10V/DIV VVIN = 3.6V C4 = 1F 40s/DIV
3587 F08b
24V
IVOUT1 500mA/DIV 0mA IL1 500mA/DIV VCAP1 10V/DIV VVOUT1 10V/DIV 15V 15V
Figure 8. VCAP3, VVOUT3, IVOUT3 and IL4 During a Short-Circuit Condition with and Without Programmed 20mA Current Limit
Choosing A Feedback Node Boost1 feedback resistor, RFB1, may be connected to the VOUT1 pin or the CAP1 pin (see Figure 9). Similarly for Boost3 in a boost voltage regulator configuration, the feedback resistor, RVFB3, may be connected to the VOUT3 pin or the CAP3 pin. Regulating the VOUT1 and VOUT3 pins eliminates the output offset resulting from the voltage drop across the output disconnect PMOS transistors.
VVIN = 3.6V C1 = 4.7F
40s/DIV
3587 F07
Figure 7. VCAP1,VVOUT1, IVOUT1 and IL1 During a Short-Circuit Event
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LT3587 APPLICATIONS INFORMATION
FLT CAP3 RVFB3 VFB3 VOUT3 IFB3 SW2 DN FB2 B1 B3 SW3 EN/SS EN/SS VIN SW1 GND CAP1 FB1 RFB1 VOUT1
LT3587
VVOUT1 +IVOUT1 * RDISC1 - 1.22V 13.8A V +I *R - 0.8V RFB3 = VOUT3 VOUT3 DISC3 14.3A RFB1 = Fault Detection and Indicator The LT3587 features fault detection on all its outputs and a fault indicator pin (FLT). The fault detection circuitry is enabled only when at least one of the channels has completed the soft-start process and is free running with full inductor current. Once the fault detection is enabled, the Fault pin pulls low when any of the feedback voltages (VFB1, VFB2 or Max(VVFB3,VIFB3)) fall below their regulation value for more than 16ms. One particularly important case is an overload or shortcircuit condition on any of the channel outputs. In this case, if the corresponding loop is unable to bring the output back into regulation within 16ms, a fault is detected and the Fault pin is pulled low. Note that the fault condition is latched. Once the Fault pin is pulled low, all the three channels are disabled. In order to enable any of the channels again, reset the part by shutting it down and then turning it on again. This is done by first forcing both the EN/SS1 and EN/SS3 pins low below 200mV and then either letting them go high again in a soft-start process or forcing them high immediately if no soft-start is desired. Figure 10 shows the waveforms when a short-circuit condition occurs at Boost1 for more than 16ms as well as the subsequent resetting of the part.
VFLT 5V/DIV ENSS1/ENSS3 5V/DIV VVOUT1 10V/DIV VNEG 10V/DIV VVOUT3 20V/DIV 100ms/DIV
3587 F10
FLT CAP3 RVFB3 VFB3 VOUT3 IFB3
B1 B3 SW3 EN/SS EN/SS
VIN
SW1 GND CAP1
LT3587 FB1 SW2 DN FB2 VOUT1
RFB1
3587 F09
Figure 9. Feedback Connection Using the VOUT and CAP Pins
However, in the case of a short-circuit fault at the VOUT pins, the LT3587 will switch continuously because the FB1 or the VFB3 pin is low. While operating in this open-loop condition, the rising voltage at the CAP pins is limited only by the protection circuit of their respective output disconnects. At the worst case, the CAP pin rises to 10V above the corresponding VOUT pin. So in the case of shortcircuit fault to ground, the voltage on the CAP pins may reach 10V. When the short-circuit condition is removed, the VOUT pins rise up to the voltage on the CAP pins, potentially exceeding the programmed output voltage until the capacitor voltages fall back into regulation. While this is harmless to the LT3587, this should be considered in the context of the external circuitry if short-circuit events are expected. Regulating the CAP pins ensures that the voltage on the VOUT pins never exceeds the set output voltage after a short-circuit event. However, this setup does not compensate for the voltage drop across the output disconnect, resulting in an output voltage that is slightly lower than the voltage set by the feedback resistor. This voltage drop is equal to the product of the output current and the on resistance of the PMOS disconnect transistor. This drop can be accounted for when using the CAP pin as the feedback node by setting the output voltage according to the following formula:
PART RESET
SHORT AT VOUT1
Figure 10. Waveforms During Fault Detection of a Short-Circuit Event
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LT3587 APPLICATIONS INFORMATION
Besides acting as a fault output indicator, the Fault pin is also an input pin. If this pin is externally forced low below 400mV, the LT3587 behaves as if a fault event has been detected and all the channels turn off. In order to turn the part back on, remove the external voltage that forces the pin low and reset the part. Figure 11 shows the waveforms when the Fault pin is externally forced low and the subsequent resetting of the part.
VFLT 5V/DIV ENSS1/ENSS3 5V/DIV VVOUT1 10V/DIV VNEG 10V/DIV VVOUT3 20V/DIV 100ms/DIV
3587 F11
Since the programmed VOUT3 current is proportional to the current through RIFB3, the LED current can be adjusted according to the following formula: IVOUT3 = (0.8V - VDAC-OUT) * 200/RIFB3 A higher DAC output voltage level results in lower LED current and hence lower overall brightness. Conversely, a lower DAC output voltage results in higher LED current and higher brightness. Note that the DAC output impedance should be low enough to be able to sink approximately 1/200 of the desired maximum LED current without any appreciable error for accurate dimming control. Note also that the maximum output current is limited by the output disconnect current limit to 110mA (typ). PWM Dimming Changing the forward current flowing in the LEDs not only changes the brightness intensity of the LEDs, it also changes the color. The chromaticity of the LEDs changes with the change in forward current. Many applications cannot tolerate any shift in the color of the LEDs. Controlling the intensity of the LEDs with a direct PWM signal allows dimming of the LEDs without changing the color. In addition, direct PWM dimming offers a wider dimming range to the user.
VVIN 2.5V TO 5V 10H
FLT FORCED LOW PART RESET
Figure 11. Waveforms When the Fault Pin is Externally Forced Low
Dimming Control For Boost3 Current Regulator as an LED Driver As shown on the front page application and the Block Diagram, one of the most common applications for the Boost3 channel when configured as a boost current regulator is a backlight LED driver. In an LED driver application, there are two different ways to implement a dimming control of the LED string. The LED current can be adjusted either by using a digital to analog converter (DAC) with a resistor RIFB3 or by using a PWM signal. Using a DAC and a Resistor For some applications, the preferred method of brightness control is using a DAC and a resistor. The Boost3 configuration for using this method is shown in Figure 12.
VIN LT3587
SW3 CAP3 VOUT3
1F
LED DRIVER
IFB3 RIFB3 8.06k
EN/SS3
DAC LTC2630
VDAC-OUT
3587 F12
Figure 12. Dimming Using a DAC and a Resistor
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LT3587 APPLICATIONS INFORMATION
VVIN 2.5V TO 5V 10H
VIN
SW3 CAP3 LT3587 VOUT3
1F
LED DRIVER 20mA
IFB3 RIFB3 8.06k
EN/SS3
Dimming the LEDs via a PWM signal essentially involves turning the LEDs on and off at the PWM frequency. The typical human eye has a sensitivity limit of ~60Hz. By increasing the PWM frequency to ~80Hz or higher, the eye will interpret that the pulsed light source is continuously on. Additionally, by modulating the duty cycle (amount of "on-time"), intensity of the LEDs is controlled. The color of the LEDs remains unchanged in this scheme since the LED current is either zero or a constant value. Figure 13 shows a partial application showing an LED driver for six white LEDs. If the voltage at the CAP3 pin is higher than 10V when the LEDs are on, direct PWM dimming method requires an external NMOS. This external NMOS is tied between the cathode of the lowest LED in the string and ground as shown in Figure 13. A Si1304 logic-level MOSFET can be used since its source is connected to ground, and it is able to withstand the open-circuit voltage at the VOUT3 pin across its drain and source. The PWM signal must be applied to the EN/SS3 pin of the LT3587 and the gate of the NMOS. The PWM signal should traverse between 0V to 2.5V, to ensure proper turn on and off of the Boost3 regulation loop and the NMOS transistor MN1. When the PWM signal goes high, the LEDs are connected to ground and a current of IVOUT3 = 160V/RIFB3 flows through the LEDs. When the PWM signal goes low, the LEDs are disconnected and turned off. The output disconnect feature and the external NMOS ensure that the LEDs quickly turn off without discharging the output capacitor. This allows the LEDs to turn on faster. Figure 14 shows the PWM dimming waveforms for the circuit in Figure 13.
2.5V 0V
PWM FREQ
MN1 Si1304BDL
3587 F13
Figure 13. Six White LEDs Driver With PWM Dimming
IVOUT3 13mA/DIV
0mA
IL4 200mA/DIV ENSS3 5V/DIV
0mA
0V
VVIN = 3.6V 6 LEDs
2ms/DIV
3587 F14
Figure 14. PWM Dimming Waveforms
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LT3587 APPLICATIONS INFORMATION
100 1kHz AVERAGE CURRENT (mA) 10
1
IDEAL MEASURED
300Hz
0.1 100Hz 0.01
1
10 DUTY CYCLE (%)
100
3587 F15
1
10 PWM DIMMING RANGE
100
3587 F16
Figure 15. Average LED Current Variation with PWM Duty Cycle at 100Hz PWM Frequency
Figure 16. Dimming Range Comparison of Three PWM Frequencies
The time it takes for the LED current to reach its programmed value sets the achievable dimming range for a given PWM frequency. Figure 15 shows the average current variation over duty cycle for a 100Hz PWM frequency with the circuit in Figure 13. Notice that at lower end of the duty cycle, the linear relation between the average LED current and the PWM duty cycle is no longer preserved. This indicates that the loop requires a fixed amount of time to reach its final current. When the duty cycle is reduced such that the amount of on time is in the order of or less than this settling time, the loop no longer has the time to regulate to its final current before it is turned off again and the initial current before settling is a larger proportion of the average current. Depending on how much linearity on the average LED current is required, the minimum LED on time is chosen based on the graphs in Figure 15. For example, for approximately 10% deviation from linearity at the lower duty cycle, the minimum on time of the LED current is approximately 320s for a 3.6V input voltage. The achievable dimming range for this application with a 100Hz PWM frequency can be determined using the following method.
Example: f = 100Hz tPERIOD = 1/f = 0.01s, tMIN-ON = 320s Dim Range = tPERIOD/tMIN-ON = 0.01s/320s 30:1 Min Duty Cycle = (tMIN-ON/tPERIOD) * 100 = 3.2% Duty Cycle Range = 100% 3.2% at 100Hz The calculations show that for a 100Hz signal the dimming range is 30 to 1. In addition, the minimum PWM duty cycle of 3.2% ensures that the LED current varies linearly with duty cycle to within 10%. Figure 16 shows the dimming range achievable for three different frequencies with a minimum on time of 320s. The dimming range can be further extended by combining this PWM method with the DAC and resistor method discussed previously. In this manner both analog dimming and PWM dimming extend the dimming range for a given application. The color of the LEDs no longer remains constant because the forward current of the LED changes with the output voltage of the DAC. For the six LED application described above, the LEDs can be dimmed first by modulating the duty cycle of the PWM signal with the DAC output at 0V. Once the minimum duty cycle is reached, the value of the DAC output voltage can be increased to further dim the LEDs. The use of both techniques together allows the average LED current for the six LED application to be varied from 20mA down to less than 1A.
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LT3587 APPLICATIONS INFORMATION
Lower Input Voltage Applications The LT3587 can be used in lower input voltage applications. The VIN supply voltage to the LT3587 must be 2.5V to 6V. However, the inductors can be run off a lower voltage. This allows the outputs to be powered off two alkaline cells. Most portable devices and systems have a 3.3V logic supply voltage which can be used to power the LT3587. The outputs can be driven straight from the battery, resulting in higher efficiency. Figure 17 shows a typical digital still camera application with CCD positive and negative supply as well as an LED driver powered by two AA cells. The battery is connected to the input inductors and the chip is powered with a 3.3V logic supply voltage.
LED DRIVER 20mA UP TO 12V RVFB3 787k (OPTIONAL) RIFB3 8.06k VFB3 VOUT3 CAP3 IFB3 FLT EN/SS1 EN/SS3
C4 1F DS3
L4 10H
L1 15H
2AA CELLS 2V TO 3.2V
SW3
SW1 CAP1
DS1 C1 4.7F CFB1 3.3pF
LT3587 RFB1 787k FB1 VIN SW2 GND FB2 VOUT1 CCD POSITIVE 12V, 10mA CFB2 6.8pF CCD NEGATIVE -8V, 20mA C7 10F
3587 F17
3.3V
2AA CELLS 2V TO 3.2V
C6 1F L2 15H
C2 2.2F
L3 15H DS2
RFB2 1M
C1: MURATA GRM21BR61E475KA12L C2: MURATA GRM188R61C225KE15D C4: MURATA GRM188R61E105KA12B C6: MURATA GRM155R61A105KE15D C7: MURATA GRM21BR71A106KE51L
CFB1: MURATA GRM1555C1H3R3BZ01D CFB2: MURATA GRM1555C1H6R3BZ01D L1, L2, L3: SUMIDA CDRH2D18/HP-150N L4: TOKO 1071AS-100M DS1, DS2, DS3: NXP PMEG2005EB
Figure 17. 2 AA Cells Providing CCD Positive and Negative Supply and a Three White Backlight LED Driver
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LT3587 APPLICATIONS INFORMATION
Board Layout Consideration As with all switching regulators, careful attention must be paid to the PCB board layout and component placement. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. In order to minimize magnetic field radiation, reduce the parasitic inductance by keeping the traces that conduct high switching currents short, wide and with minimal overall loop area. These are typically the traces associated with the switches. Figure 18 outlines the critical paths.
VVIN L1 DS1 VVIN SW1 CAP1 C6 Q1 LT3587 GND C1 C6 Q3 L4 DS3
The voltage signals of the SW1, SW2 and SW3 pins have rise and fall times of a few ns. Minimize the length and area of all traces connected to the SW1, SW2 and SW3 pins to reduce capacitive coupling between these fast nodes and other circuitry. In particular, keep all the traces of the feedback voltage pins (FB1, FB2, VFB3 and IFB3) away from the switching node. Always use a ground plane under the switching regulator to minimize interplane coupling. Finally, place as much of the output capacitors of each channel close to their respective CAP pins. Recommended component placement is shown in Figure 19.
VVIN
L2
C2
L3 VNEG
SW3 CAP3 LT3587 GND C4 C6
SW2 DS2
Q2
LT3587 GND
C7
3587 F18
Figure 18. High Current Paths
VIN VNEG CAP3
C6 (OPT) C7 L4 RFB2 L3 CFB2 C2 DS2 DS1 CFB1 L2 L1 C1 U1 RFB1 DS3 C5 C6 C3 C4 RIFB3
VOUT3
VIN
VOUT1
C6 (OPT)
C6 (OPT)
3587 F19
VIN
CAP1
Figure 19. Recommended Component Placement
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LT3587 TYPICAL APPLICATIONS
Li-Ion Powered Supply for CCD Imager and Five White Backlight LEDs
LED DRIVER 20mA UP TO 24V
RVFB3 1.65M (OPTIONAL)
C4 2.2F DS3
L4 10H
L1 15H
VVIN 2.5V TO 6V C6 1F SW1
VFB3 VOUT3 CAP3 RIFB3 8.06k IFB3
SW3
VIN
DS1 C1 10F
CAP1 CFB1 2.7pF
LT3587 FLT EN/SS1 EN/SS3 C3 100nF SW2 C5 100nF C2 2.2F GND FB2 FB1 VOUT1
RFB1 1M CCD POSITIVE 15V, 50mA CFB2 6.8pF CCD NEGATIVE -8V, 100mA C7 22F
3587 TA02
VVIN 2.5V TO 6V
L2 15H
L3 15H DS2
RFB2 1M
C1: MURATA GRM21BR61C106KE15L C2: MURATA GRM188R61C225KE15D C3, C5: MURATA GRM033R60J104KE19D C4: MURATA GRM21BR71E225KA73L C6: MURATA GRM155R61A105KE15D C7: TAIYO YUDEN LMK212BJ226MG-T
CFB1: MURATA GRM1555C1H2R7BZ01D CFB2: MURATA GRM1555C1H6R8BZ01D L1, L2, L3: SUMIDA CDRH2D18/HP-150N L4: TOKO 1071AS-100M DS1, DS2, DS3: IR IR05H40CSPTR
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LT3587 TYPICAL APPLICATIONS
Driver For a CCD Imager and an OLED Display Panel with Soft-Start
OLED DRIVER 16V, 20mA RVFB3 1.07M RIFB3 7.15k (OPTIONAL) VFB3 IFB3 LT3587 FLT EN/SS1 EN/SS3 C3 100nF C5 100nF VVIN 2.5V TO 6V L2 15H SW2 GND FB2 FB1 VOUT1 VOUT3 CAP3 C4 1F VVIN 2.5V TO 6V L4 10H
DS3
L1 15H
C6 1F SW1 CAP1 CFB1 2.7pF RFB1 1M C1 DS1 4.7F
SW3
VIN
CCD POSITIVE 15V, 50mA
C2 2.2F
D3
RFB2 1M C7 22F
CFB2 6.8pF
3587 TA03
CCD NEGATIVE -8V, 100mA
DS2
C1: TAIYO YUDEN TMK212BJ475KG-T C2: TAIYO YUDEN EMK107BJ225KA-T C3, C5: TAIYO YUDEN JMK063BJ104KP-F C4: TAIYO YUDEN GMK107BJ105KA-T C6: TAIYO YUDEN LMK105BJ105KV-F
C7: TAIYO YUDEN LMK212BJ226MG-T CFB1: TAIYO YUDEN EMK105SK2R7JW-F CFB2: TAIYO YUDEN EMK105SH6R8JW-F L1, L2: SUMIDA CDRH2D18/HP-150N L4: TOKO 1071AS-100M DS1, DS2, DS3, D3: NXP PMEG2005EB
Extending the High Voltage Range and the Number of Independently Controlled Regulated Outputs
D8 24V GND SHDN SUPPLY 5: 40V, 5mA OUT R6 3.24M R7 100k LT3014 ADJ SHDN GND SUPPLY 3: 25V, 1mA TO 10mA RVFB3 1.74M RIFB3 10.7k VFB3 IFB3 FLT EN/SS1 EN/SS3 LT3587 RFB1 1.21M VOUT3 CAP3 SUPPLY 1 C4 2.2F DS3 SUPPLY 2 D10 VVIN 3V TO 6V IN C12 2.2F D6 D4 C9 2.2F D7 C11 2.2F C8 2.2F D5 IN ADJ R6 1.21M SUPPLY 4: -40V, 5mA R5 100k C10 1F
LT1964 OUT
C13 1F
L4 15H
L1 15H DS1
C6 1F C1 10F
SW3
VIN
SW1 CAP1 CFB1 2.2pF
FB1 SW2 GND C2 2.2F FB2 RFB2 2.26M VOUT1
C3 100nF
C5 100nF VVIN 3V TO 6V
SUPPLY 1: 18V, 50mA CFB2 2.7pF SUPPLY 2: -18V, 50mA
3587 TA04
L2 22H
D3
D9
DS2
C7 22F
C1: MURATA GRM31CR71E106KA12L C2, C8, C11: MURATA GCM21BR71E225KA73L C3, C5: MURATA GRM033R60J104KE19D C4: MURATA GRM21BR71E225KA73L C6: MURATA GRM155R61A105KE15D C7: MURATA GRM32ER61E226KE15L C9, C12: MURATA GRM31CR71H225KA55L C10, C13: MURATA GRM21BR71H105KA12L
CFB1: MURATA GRM1555C1H2R2BZ01D CFB2: MURATA GRM1555C1H2R7BZ01D L1, L4: COILCRAFT LPS4018-153 L2, L3: COILCRAFT LPS4018-223 DS1, DS2, DS3, D3, D4, D5, D6, D7, D9, D10: IR IR05H40CSPTR D8: DIODES INC DDZ9709T
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22
LT3587 PACKAGE DESCRIPTION
UD Package 20-Lead Plastic QFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1720 Rev A)
0.70 0.05 3.50 0.05 (4 SIDES)
1.65 0.05
2.10 0.05
PACKAGE OUTLINE 0.20 0.05 0.40 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED BOTTOM VIEW--EXPOSED PAD PIN 1 NOTCH R = 0.20 TYP OR 0.25 x 45 CHAMFER 19 20 0.40 0.10 1 2 1.65 0.10 (4-SIDES)
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (NOTE 6)
0.75 0.05 R = 0.05 TYP
R = 0.115 TYP
(UD20) QFN 0306 REV A
0.200 REF 0.00 - 0.05 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
0.20 0.05 0.40 BSC
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3587 TYPICAL APPLICATION
General Purpose High Voltage Supplies Generator
SUPPLY 3: 25V SUPPLY WITH SAFETY CURRENT LIMIT AT 25mA C4 2.2F RVFB3 1.74M RIFB3 6.34k VFB3 VOUT3 IFB3 LT3587 FLT EN/SS1 EN/SS3 C3 100nF C5 100nF VVIN 2.5V TO 6V L2 22H FB1 SW2 GND FB2 RFB2 2M VOUT1 RFB1 1M C1: MURATA GRM31CR71E106KA12L C2: MURATA GRM21BR71E225KA73L C3, C5: MURATA GRM033R60J104KE19D C4: MURATA GRM21BR71E225KA73L C6: MURATA GRM155R61A105KE15D C7: MURATA GRM32ER61E226KE15L CFB1: MURATA GRM1555C1H2R7BZ01D CFB2: MURATA GRM1555C1H2R2BZ01D L1: COILCRAFT LPS4018-153 L2, L3: COILCRAFT LPS4018-223 L4: TOKO 1071AS-100M DS1, DS2, DS3, DS4: IR IR05H40CSPTR CAP3
DS3
L4 10H
L1 15H
VVIN 2.5V TO 6V C6 1F SW1 CAP1 CFB1 2.7pF DS1 C1 10F
SW3
VIN
SUPPLY 1: 15V, 50mA
C2 2.2F
L3 22H DS2
CFB2 2.2pF SUPPLY 2: -16V, 50mA
3587 TA05
DS4
C7 22F
RELATED PARTS
PART NUMBER LT1944 LT1945 LT3463/LT3463A DESCRIPTION Dual Output, Boost/Inverter, 350mA ISW, High Efficiency Boost-Inverting DC/DC Converter Dual Output, Boost/Inverter, 350mA ISW, High Efficiency Boost-Inverting DC/DC Converter Dual Output, Boost/Inverter, 250mA ISW, Constant OffTime, High Efficiency Step-Up DC/DC Converter with Integrated Schottkys Dual Constant Current, 2MHz, High Efficiency White LED Boost Regulator with Integrated Schottky Diode Dual Output, Boost/Inverter, 1.3A ISW, 1.2MHz, High Efficiency Boost-Inverting DC/DC Converter Dual Output, Boost/Inverter, 400mA ISW, 1.2MHz, High Efficiency Boost-Inverting DC/DC Converter COMMENTS VIN(MIN) = 1.2V, VIN(MAX) = 15V, VOUT(MAX) = 34V, IQ = 20, ISD < 1A, MSOP-10 Package VIN(MIN) = 1.2V, VIN(MAX) = 15V, VOUT(MAX) = 34V, IQ = 20, ISD < 1A, MSOP-10 Package VIN(MIN) = 2.3V, VIN(MAX) = 15V, VOUT(MAX) = 40V, IQ = 40A, ISD < 1A, 3 x 3 DFN-10 Package VIN(MIN) = 2.7V, VIN(MAX) = 24V, VOUT(MAX) = 40V, IQ = 5mA, ISD < 16A, 3 x 3 DFN-10 Package VIN(MIN) = 2.4V, VIN(MAX) = 16V, VOUT(MAX) = 40V, IQ = 2.5mA, ISD < 1A, 3 x 3 DFN-10 Package VIN(MIN) = 2.2V, VIN(MAX) = 16V, VOUT(MAX) = 34V IQ = 2.8mA, ISD < 1A, 3 x 3 DFN-10 Package
LT3466/LT3466-1 LT3471 LT3472/LT3472A LT3473/LT3473A LT3494/LT3494A LT3497 LT3580
40V, 1A, 1.2MHz Micropower Low Noise Boost Converter VIN(MIN) = 2.2V, VIN(MAX) = 16V, VOUT(MAX) = 36V, IQ = 150A, ISD < 1A, with Output Disconnect 3 x 3 DFN-12 Package 40V, 180mA/350mA Micropower Low Noise Boost Converter with Output Disconnect Dual 2.3MHz, Full Function LED Driver with Integrated Schottkys and 250:1 True Color PWM Dimming 40V, 2A, 2.5MHz Boost/Inverter DC/DC Converter VIN(MIN) = 2.3V, VIN(MAX) = 16V, VOUT(MAX) = 40V, IQ = 65A, ISD < 1A, 3 x 2 DFN-8 Package VIN(MIN) = 2.5V, VIN(MAX) = 10V, VOUT(MAX) = 32V, IQ = 6mA, ISD < 12A, 2 x 3 DFN-10 Package VIN: 2.5V to 32V, VOUT(MAX) = 40V, IQ = 1mA, ISD < 1A, MSOP-8E, 3mm x 3mm DFN-8 Packages
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24 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0109 REV C * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2008

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